RF-MEMS Switches for Reconfigurable

RF-MEMS Switches for Reconfigurable
Integrated Circuits

Abstract — This paper deals with a relatively new area of radio-frequency (RF) technology based on microelectromechanical systems (MEMS). RF MEMS provides a class of new devices and components which display superior high-frequency performance relative to conventional (usually semiconductor) devices, and which enable new system capabilities. In addition, MEMS devices are designed and fabricated by techniques similar to those of very large-scale integration, and can be manufactured by traditional batch-processing methods. In this paper, the only device addressed is the electrostatic microswitch—perhaps the paradigm RF-MEMS device. Through its superior performance characteristics, the microswitch is being developed in a number of existing circuits and systems, including radio front-ends, capacitor banks, and time-delay networks. The superior performance combined with ultra-low-power dissipation and large-scale integration should enable new system functionality as well. Two possibilities addressed here are quasi-optical beam steering and electrically reconfigurable antennas.

 I.  INTRODUCTION

The 1990's have brought a profound change in radio-frequency (RF) technology driven largely by economic Tand geopolitical events. On one hand, the wind-down of the cold war has reduced the need for advanced RF systems, particularly sensors; on the other hand, the dawning of the information age has created a heightened interest and world-wide market for  communications  systems and networking of voice and data alike.The transition of RF technology from one era to the other has been both challenging and opportunistic. For the RF systems engineers, it has meant a shift of thinking from large centralized systems to smaller distributed systems. Along with this shift has come a change from long-range systems, having large RF transmit power, to shorter range systems, having relatively modest RF power. In many cases, the new smaller  systems must be mobile or hand-held. The paradigm for these new systems is the cellular wireless network consisting of a single powerful base station feeding a local cell of hand sets acting like individual terminals or nodes of the network. The popular digital cellular and personal communications service (PCS) bands around 0.9 and 1.9 GHz, respectively, comprise much of the frequency spectrum being used for cellular purposes.
 

For technology engineers, the transition has been no less challenging. The premium devices and components formerly required to construct powerful centralized systems are no longer required or can no longer be afforded in many new distributed systems coming on line today. Instead, there is an emphasis on more affordable and integrable technology, which allows a greater degree of RF functionality per unit volume, even if at a lower level of performance than obtained with the former technologies. This has spawned widespread research and development of silicon-based RF integrated circuits (RFIC's), including deep-submicrometer Si CMOS and SiGe heterojunction bipolar transistors (HBT's). Taking advantage of the inherent manufacturability of Si very large-scale integration (VLSI), RFIC technology has found unique circuit and subsystem architectures well outside the traditional digital design. One example of this is the "RF system-on-a-chip," such as the family of integrated circuits (IC's) now commercially available for global positioning receivers.

This paper deals with another technology that has emerged in recent years with a comparable level of interest and more rapid development than RFIC's. The technology is the design and fabrication of microelectromechanical systems (MEMS) for RF circuits (RF MEMS). In some ways, MEMS represents the new revolution in microelectronics. It is similar to VLSI circuits in that it allows the execution of complex functions on a size scale orders of magnitude lower and at far less power than discrete circuits. However, MEMS enables this miniaturization on a class of sensors and transducers that traditionally were constructed on the model of a large, often cumbersome transducer or sensor coupled to a highly integrated VLSI readout circuit or processor. A good example of this is the MEMS accelerometer, now one of the largest single MEMS application through its incorporation in air bags [1]. At the same time, MEMS leverages VLSI through the use of common design and batch processing methodologies and tools. It is this commonality with VLSI that has been credited to a large extent for the rapid dissemination of MEMS into the commercial marketplace.

It is important to realize up front that RF MEMS does not necessarily imply that the micromechanical system is operating at RF frequencies. As will be discussed briefly, in the largest class of RF MEMS devices and components, the microelectromechanical operation is used simply for the actuation or adjustment of a separate RF device or component, such as a variable capacitor. In many of these devices, a key advantage of the MEMS devices compared to traditional semiconductor devices is electromechanical isolation. By this, we mean that the  RF circuit does not leak or couple significantly to the actuation circuit. A second advantage is power consumption. Many of the RF MEMS devices under development carry out electromechanical coupling electrostatically through air (or vacuum). Hence, the power consumption comes from dynamic current flowing to the MEMS only when actuation is occurring. However, the implementation of RF MEMS does not come with  impunity. Due to the mechanical actuation, they are inherently slower than electronic switches. The electromechanical actuation time is typically many microseconds or greater, which is substantially longer than typical electrical time constants in semiconductor devices. In addition, RF MEMS devices can exhibit the phenomenon of "stiction," whereby parts of the device can bonded together upon physical contact. Each of these issues will be discussed further.

II.  MEMS AND  MICROMACHINING

 According to a recent definition, a MEMS is a miniature device or an array of devices combining electrical and mechanical components and fabricated with IC batch-processing techniques [2]. Critical to this definition is that MEMS has both device and fabrication aspects. There are several MEMS fabrication techniques currently in widespread use, including bulk micromachining, surface micromachining, fusion bonding, and LIGA, which is a composite fabrication procedure of lithography, electroforming, and molding. The most important technique for RF MEMS is surface micromachining. In short, surface micromachining consists of the deposition and lithographic patterning of various thin films, usually on Si substrates. Generally, the intent is to make one or more of the ("release") films freestanding over a selected part of the substrate, thereby able to undergo the mechanical motion or actuation  characteristic of all MEMS. This is done by depositing a "sacrificial" film (or films) below the released one(s), which is removed in the last steps of the process by selective etchants. The  variety of materials for the release and sacrificial layers is great, including many metals (Au, Al, etc.), ceramics (SiO and Si N ), and plastics (photoresist, polymethyl methacrylate (PMMA), etc.). Depending on the details of the MEMS process and the other materials in the thin-film stack, the release and sacrificial layers can be deposited by evaporation, sputtering, electrodeposition, or other methods.

Surface micromachining has been used for a long time, dating back to MEMS work of the 1960's at Westinghouse. A breakthrough in surface micromachining has come in the form of dry etching, particularly reactive-ion etching (RIE). By mixing reactive chemicals in a plasma discharge  and adding a semiconductor wafer with thin films deposited on top, select materials on the surface can be etched away at useful high rates and with high levels of material selectivity. For example, chlorine-bearing compounds in a high-density plasma can yield nearly isotropic silicon etching with a selectivity of silicon-to-SiO  of better than 100 : 1. By the same token, low-pressure plasma etching [e.g., inductively coupled plasma (ICP)] allows independent control of the ion density and energy.

Neyker Stewart Zambrano

CRF

http://my.ece.ucsb.edu/yorklab/Useful%20Stuff/References/Papers/Brown%20RF%20MEMS%20switches%20MTT%20Nov98.pdf

Monolithic Microwave Integrated Circuits

Monolithic Microwave Integrated Circuits

Monolithic microwave integrated circuits (MMICs) and similar devices are used in a wide variety of receivers. These devices may be very wideband or relatively narrowband. Very wideband amplifiers have a bandpass (frequency response) of several hundred megahertz or more, typically ranging from sub-VLF to the low end of the microwave spectrum. An example might be a range of 100 kHz to 1000 MHz (i.e.,  1 GHz), although somewhat narrower ranges  are more common. These circuits have a variety of practical uses: receiver preamplifiers, signal generator output amplifiers, buffer amplifiers in RF instrument circuits, cable television line amplifiers, and many others in communications and instrumentation.

One reason why very wideband amplifiers are rarer than narrowband amplifier circuits is that they were difficult to design and build until the advent of monolithic microwave integrated circuit devices. Several factors contribute to the difficulty of designing and building very wideband amplifiers. For example, too many stray capacitances and inductances are in a typical circuit layout, and these form resonances and filters that distort the frequency response characteristic. Also, circuit resistances combine with the capacitances to effectively form low-pass filters that roll off the frequency response at higher frequencies, sometimes drastically.  If the RC phase shift of the circuit resistances and capacitances is 180º at a frequency where the amplifier gain is ≥1 (and in very wideband circuits that is likely) and the amplifier is an inverting type (producing an inherent 180º phase shift), then the total end-to-end phase shift is 360º—the criteria for self-oscillation.

If you have ever tried to build a very wideband amplifier, it likely was a very frustrating experience. Until now. New, low-cost devices, called silicon MMICs, make it possible to design and build amplifiers that cover the spectrum from near DC to about 2000 MHz,  using seven or fewer components. These devices offer 13–30 dB of gain and produce output power levels up to 40 mW (+16 dBm). Noise figures range from 3.5 to 7 dB. In this chapter, we use the MAR-X series of MMICs by Mini-Circuits Laboratories as representative.

The only connections are RF input, RF output, and two ground connections. The use of dual ground connections distributes the grounding, reducing overall inductance and thereby improving the ground connection. Direct current power is applied to the output terminal  through an external network. But more on that shortly.

Although an IC, the device looks very much like a small UHF/microwave transistor. The body  is made of plastic and the leads are wide metal strips (rather than wire) to reduce the stray inductance that narrower wire leads would exhibit. These devices are small enough that handling can be difficult; I found that hand forceps (tweezers) were necessary to position the device on a prototype printed circuit board. A magnifying glass or jeweler's eye loupe is not out of order for those with poor close-in eyesight. A color dot and a beveled tip on one lead are the  keys that identify pin 1 (the  RF input connection). When viewed from above, pin numbering  (1,  2,  3,  4) proceeds counter-clockwise from the keyed pin.

INTERNAL CIRCUITRY

The MAR-X series of devices inherently matches 50 Ω input and output impedances without external impedance transformation circuitry, making it an excellent choice for general RF applications. These devices are silicon bipolar monolithic ICs in a two-transistor Darlington amplifier configuration. Because of the Darlington connection, the MAR-X devices act like transistors with very high gain. Because the transistors are biased internal to the MAR-X package, the overall gains typically are 13–33 dB, depending on the device selected and operating frequency. No external bias or emitter bias resistors are needed, although a collector load resistor to V+ is used.

Neyker Stewart Zambrano

CRF

http://www.download-it.org/free_files/9399a76e85adfa8e64b99aba008e281a-Pages%20from%20Chapter%2013..pdf  

RF/MICROWAVE GLOSSARY

RF/Microwave Glossary

Albaloy: A plating finish comprised primarily of copper, tin and zinc which provides good electrical performance, but unlike silver, albaloy is highly resistant to tarnish.  Being non-magnetic, it also provides excellent passive intermodulation (PIM) performance comparable to silver.

Amplitude Balance: The maximum peak-to-peak amplitude difference (in dB) between the output ports of a power divider or hybrid coupler over the specified frequency range.

Attenuation Accuracy: The amount of variation in magnitude from the nominal value across the entire frequency band.

Attenuator: A passive device or network that absorbs part of the input signal and transmits the remainder with minimal distortion. Attenuators are used to extend the dynamic range of devices such as power meters and amplifiers, reduce signal levels to detectors, match circuits and are used daily in lab applications to aid in product design. Attenuators are also used to balance out transmission lines that otherwise would have unequal signal levels.

Base Station: A fixed transmitter/receiver with which a mobile radio transceiver establishes a connection link to gain access to the public-switched telephone network.

Bias Tees: A passive device used in applications to inject/remove DC voltages in RF circuits without affecting the RF signal through the main transmission path.  Ideal for remote powering of bi-directional amplifiers (BDAs), repeaters and tower top amplifiers (TTAs) by BTS control modules. 
Circulator: A three-port ferromagnetic passive device used to control the direction of signal flow in an RF circuit.

Coaxial: A transmission line in which one conductor completely surrounds the other, the two being coaxial and separated by a continuous dielectric such as air or PTFE.

CW – (Continuous Wave): Signal of constant amplitude.  Used to differentiate between the performance of a microwave component for continuous power level vs. pulsed signals.

dB – (Decibel): A unit of gain equal to ten times the common logarithm of the ratio of two power levels or 20 times the common logarithm of the ratio between two voltages.

dBc: Decibel related to the signal of a carrier.  Passive intermodulation distortion is typically stated in dBc which takes into consideration the 43 dBm carrier tones.

dBm: Decibels related to 1mW – the standard unit of power level used in the microwave industry.  Example: 0 dBm = 1mw,  +10 dBm = 10mw,  +20dBm = 100mw, etc.

DC Block: An in-line device primarily used in applications to block DC voltages in RF circuits without affecting the RF signal through the main transmission path.  The three basic types are:
Inner – Blocks DC voltages on inner conductor only
Outer – Blocks DC voltages on outer conductor only
Inner/Outer – Blocks DC voltages on both conductors

Directional Coupler: A passive device used for sampling incident and reflected microwave power conveniently and accurately with minimal disturbance to the transmission line. Some general applications for directional couplers include line monitoring, power measurements and load source isolators.

Directivity: A measurement of the desired signal strength to the undesired signal strength.  Determined by taking the value of isolation and subtracting the specified coupling (including all variations).  Directivity is a measure of how good the couplers performance is (similar to the Q factor of a coil).

EMI – (Electromagnetic Interference):  Unintentional interfering signals generated within or external to electronic equipment.  Typical sources could be power line transients and electromechanical switching equipment.

Frequency Range: The minimum and maximum frequencies between which the specified component will meet all guaranteed specification.

Frequency Sensitivity: The maximum peak-to-peak variation in coupling (in dB) of a directional or hybrid coupler over the specified frequency range.  Also referred to as "flatness".

GHz - (Gigahertz): A unit of frequency measure equal to 1000 MHz (Megahertz) or a billion hertz.

Hybrid Coupler: A passive four-port device that is used either to equally split an input signal with a resultant 90° phase shift between output signals or to combine two signals while maintaining high isolation between them.

Impedance: Resistance to alternating current.  Most RF and microwave systems are designed to operate with a characteristic impedance of 50 ohms.

Input VSWR: Minimum voltage standing wave ratio of a power divider at the input (sum) port over the specified frequency range with all other ports terminated in 50 ohm loads.

Insertion Loss: The change in load power due to the insertion of a particular device into a transmission system.

Iridite: A chemical film (typically clear or yellow in color) which provides a barrier medium to prevent corrosion on aluminum surfaces and enhance adhesion of subsequent coatings such as paints and primers.

Isolation:  A unit of measure (in dB) that states the separation of signal levels on adjacent ports of a device.  The greater the isolation value, less interference from a signal on one port is present at the other.

Isolator:  A two-port ferromagnetic passive device which is used to control the direction of signal flow and utilizes an internal resistor.  Typically used to protect other RF components from excessive signal reflection.

MHz - (Megahertz): A unit of frequency measure equal to 1000 kHz (Kilohertz) or a million hertz.

Microstrip – (Microstripline): A transmission line consisting of a metalized strip and solid ground plane metallization separated by a thin, solid dielectric.  Microstrip is a popular material above 400 MHz and below 6 GHz because it permits accurate fabrication of transmission lines on ceramic or PC board substrates.  Higher frequencies or broadband devices tend to favor stripline technology.

MTBF – (Mean Time Between Failure): The mean (average) time between failures of a component and is often attributed to the "useful life" of the materials used to assemble the device.  MTBF assumes that the component can be "renewed" or fixed after each failure and returned to service immediately after failure.

Non-Coherent Signals: The limiting factor for most Wilkinson power dividers used as combiners is power dissipation. When input signals are out of phase, non-coherent or have amplitude unbalance this causes a cancellation across the isolation resistors resulting in power dissipation.  Since these devices are most commonly used as dividers, typical industry designs utilize low power alumina surface mount resistor chips on a thermally insulative circuit board. However, maximum input for combining non-coherent signals on adjacent ports is:

(Rated input power of divider * 5%) / "N" # of input channelsIf the rated power is exceeded, the chip resistors will heat up and degrade resulting in loss of port-to-port isolation and VSWR.

Output VSWR: Minimum voltage standing wave ratio of a power divider at any output port over the specified frequency range with all other ports terminated in 50 ohm loads.

Passivation: The formation of an insulated layer directly over a metal to protect the surface from contaminants, moisture or particles.

Phase Balance: The maximum peak-to-peak phase difference (in degrees) between the output ports of a power divider over the specified frequency range.

PIM (Passive Intermodulation): Passive Intermodulation (PIM) occurs when two or more signals are present in a passive device (cable, connector, coupler, etc.) that exhibits a nonlinear response. The nonlinearity is typically caused by dissimilar metals or dirty/loose interconnects.   Nonlinearity is typically not troublesome at low input signal levels, but if PIM is generated from a high power transmitter path to an adjacent receiver channel, desensitization will occur.   A common PIM specification is typically -110 dBc or greater.

Power (Average): The maximum amount of mean (average) power of a modulated/pulsed signal a given component can dissipate at ambient temperature without degradation in performance.

Power (Peak): Instantaneous power a given component can dissipate for a percentage of the duty cycle (typically 2%) without degradation in performance.

PTFE (PolyTetraFluoroEthylene): Used as an insulator in RF and microwave coaxial connectors because of its low & stable dielectric constant and loss factor over a wide temperature and frequency range.

Reactive Splitter:  A broadband passive network that equally divides power applied to the input ports between any particular number of output ports without substantially affecting the phase relationship or causing distortion.  Reactive splitters differ from Wilkinson power dividers as they provide no isolation between adjacent ports.  Therefore, power entering any output of a reactive splitter will divide evenly between the adjacent and input ports.

Return Loss: When expressed in dB is the ratio of reflected power to incident power.  It is a measure of the amount of reflected power on a transmission line when it is terminated or connected to any passive or active device.  Once it is measured, it can be converted by equation to reflection coefficient which can be converted to VSWR.

RF – (Radio Frequency): Generally referring to any frequency at which the radiation of electromagnetic energy is possible typically above 50 MHz.  Above 1000 MHz and up is considered microwave.

RF Leakage: The amount of energy which "leaks" or radiates from a connector and/or device.  Typically tested at one frequency and expressed in dB.  Very large negative values indicate that the device does not radiate much energy.

RoHS: (Restriction of Hazardous Substances) Directive adopted by the European Union in February 2003 with the specified limits for the following elements in the manufacture of various types of electronic and electrical equipment:
Lead (Pb) < 0.1%
Mercury (Hg) < 0.1%
Cadmium (Cd) < 0.01%
Hexavalent Chromium (CrVI) < 0.1%
Polybrominated Biphenyls (PBB) < 0.1%
Polybrominated Diphenyl Esters (PBDE) < 0.1%

Stripline: A transmission line consisting of a conductor above or between extended conducting surfaces.  Higher frequencies or broadband devices tend to favor stripline technology.

Termination (RF Loads): Used at the end of a transmission line designed to absorb RF power with very little reflection, effectively terminating the line or port in its characteristic impedance. Terminations are used in a wide variety of measurement systems; any port of a multi-port microwave device that is not involved in the measurement should be terminated in its characteristic impedance in order to ensure an accurate measurement.

Temperature: The minimum and maximum ambient temperatures a given component can operate at and still meet all guaranteed specifications unless otherwise noted.

Torque:  Recommended mating torque for industry standard connectors:
SMA - 7 to 10 in-lbs
Type-N – 12 to 15 in-lbs
TNC – 12 to 15 in-lbs
7/16 DIN – 220 to 300 in-lbs

Transmission Line: The conductive connections between circuit elements which carry signal power.  Wire, coaxial cable, microstrip and stripline traces and waveguide are common examples.

VSWR – (Voltage Standing Wave Ratio): The ratio of the incident signal compared to the reflected signal in a transmission line.  VSWR cannot be directly measured, so a return loss measurement (expressed in dB) is taken of reflected power to incident power.  Once it is measured, it can be converted by equation to reflection coefficient which can be converted to VSWR.

Wilkinson Power Divider: A passive device that equally splits an input signal to each output or combines signals to a common port.  Wilkinson power divider differ from reactive splitters as the output ports are isolated, so signals entering one of the output ports will not interfere with signals on the adjacent port.  The limiting factor for Wilkinson power dividers used as combiners is power dissipation. When input signals are out of phase, non-coherent or have amplitude unbalance this causes a cancellation across the isolation resistors resulting in power dissipation.

Neyker Stewart Zambrano

CRF

http://www.e-meca.com/tech_papers/glossary.php

TRFIC and Microwave Circuits

Frequency divider design strategies

Typically, in frequency divider design, the trade offs are around the maximum operating frequency, power consumption, number of transistors needed and flexibility. Depending on the specific application the frequency divider is used, analog or digital approaches may be adopted. This paper will cover the fundamentals of both approaches.

The frequency divider is an important building block in today's RFIC and microwave circuits because it is an integral part of the phase-locked loop (PLL) circuit. In a typical PLL loop, the output of the voltage-controlled oscillator (VCO) is divided down by the frequency divider to a frequency the temperature-compensated crystal oscillator (TCXO) operates (typically from 10 MHz to 30 MHz). The divided signal and TCXO are fed into the phase detector for comparison. The output phase difference is used to adjust the VCO output frequency. The frequency divider is also widely used to generate a precision I/Q signal if the input signal has a 50% duty cycle, for the modern in-phase and quadrature (I/Q) modulator or demodulator. For the signal with duty cycle other than 50%, an additional divide- by-2 can be used to generate the 50% duty cycle. Compared with the traditional resistor and capacitor (RC) quadrature generation, the frequency divider approach is easier to implement, is lower power and offers smaller phase imbalance.

Digital logic approaches

Since the PLL is part of the RF circuit, one would think the frequency divider should be analyzed by the analog circuit theory. But it turns out the most widely used approach is based on the digital logic gate. The RF analog engineer typically puts on the digital logic hat for a minute when analyzing the frequency divider. It will be much easier to understand and analyze the digital approach in the digital domain and the analog approach in the analog domain. Within the digital domain, the design strategy can be further divided into two categories: static logic and dynamic logic.
The static implementation is the most popular approach. The memory cell is a true bistable circuit, unlike the parasitic capacitor used in the dynamic approach. One standard design is the divider by 2 cell. In today's design software, it is treated as a standard digital cell. There are many names for circuits, such as a Johnson divider, toggle switch, complementary metal oxide semiconductor (CMOS) prescaler, emitter couple logic (ECL) and a source couple logic (SCL). When it is implemented in CMOS, it is called SCL. When it is implemented in a bipolar process, it is logically called ECL. For this illustration, CMOS implementation (SCL) is used.
The first look at this design might be intimidating for an RF design engineer. It doesn't look like an RF circuit with the exception of the differential amplifier and cross-coupled negative channel field effect transistor (nFET) of a VCO. Treating it as a pure RF circuit will make it much harder to understand. However, once analyzed in the digital domain, the answer will be clear. Before the detailed discussion of this cback to the input port D. The same clock is used to drive both level-triggered DFF with opposite logic. The reason for the inverter is to make an edge-triggered DFF out of the two level-triggered DFF. The first DFF is commonly called the master DFF and the second one is normally referred to as the slave DFF. Either master DFF or slave DFF is activated in each clock cycle, not both DFF at the same timeircuit, some of the fundamental digital logic theory should be reviewed.
The theory of operation is straightforward. It is essentially an edge-triggered master/slave D flip-flop (DFF). The inverted output is fed back to the input port D. The same clock is used to drive both level- triggered DFF with opposite logic. The reason for the inverter is to make an edge-triggered DFF out of the two level-triggered DFF. The first DFF is commonly called the master DFF and the second one is normally referred to as the slave DFF. Either master DFF or slave DFF is activated in each clock cycle, not both DFF at the same time (because of the inverter between them), each positive input clock cycle is loaded into the DFF. On the next cycle, inverted output again is fed back to the input, which causes the output to toggle. It is why toggle DFF is a more descriptive name for this circuit. The same event repeats for every two input clock cycle. Thus, output frequency is half of the input frequency.
a typical DFF implementation is comprised of cross-coupled NOR gates. Without the inverter, it is recognizable as the familiar SR latch. The next logic step is to take a look at how the SR latch logic is implemented on the transistor level.
Transistor Q2, Q5, Q7 and Q9 form the first NOR gate while Q3, Q6, Q8 and Q10 complete the second NOR gate. It should be obvious that Q2 and Q5 are the pull-down network (PDN) for passing logic 1s and Q7 and Q9 are the pull-up network (PUN) for passing logic 0s. Similarly, the pairs (Q3, Q6) and (Q10, Q8) are the counterparts for the second NOR gate, the circuit is the same circuit used for the edge-triggered DFF SCL implementation with minor modifications. Q2, Q5, Q3 and Q6 are still the same circuit elements rearranged slightly. Q7 and Q8 are still the active loads. Q1 and Q4 act as the input buffer. They also provide the enable function. As discussed earlier, in edge-triggered master-slave DFF, only one DFF is activated at the time. The inverter can be easily implemented by swapping the position of the "+RF" and "-RF" when feeding different DFFs.

Dynamic DFF implementation

In dynamic DFF implementation, there is no dedicated bistable circuitry. The parasitic cap between the node acts as the storage element. It is typically called Clocked CMOS (C2MOS).
This circuit uses a far less number of transistors. The theory ofoperation is simple. Transistors Q1 to Q4 complete essentially a tri-state inverter (INV1). Transistors Q5 to Q9 (INV2) form another inverter. The capacitor in the middle is a model for the parasitic capacitance between the gate. The capacitor's responsibility is to store the signal. Q9 and Q10 make a simple CMOS inverter to complete the feedback path needed for the toggle latch. In the positive clock cycle (RF+), INV1 is on and INV2 is off, so the signal is clocked into the storage capacitor. In the negative clock cycle (RF-), INV1 is off and INV2 is on, the signal is clocked out. There are many different flavors using a similar theory of operation.
The circuit requires the complementary clock input (RF+ and RF-). Sometimes, it is more desirable to drive the frequency divider single ended. The type of logic is called true single-phase clocked (TSPC) logic. It is built on the basic C2MOS. It eliminates the differential drive requirement at the expense of more transistors.
Transistors Q1 to Q6 form the master latches. Transistors Q7 to Q12 form the slave latches. Q13 and Q14 are simply the inverter to complete the feedback path needed in the toggle latch. The master latch is sometimes called double NC2MOS because two NFETs are needed in the PDN. Similarly, the slave latch is sometimes referred to as the double PC2MOS because two PFETs are needed in the PUN. When the clock is high, the master latch is activated while the slave latch is off. The opposite is true when the clock is low.
Comparing the static and dynamic implementation, static logic is faster and more reliable. The dynamic logic uses a far fewer number of transistors and is easier to implement. However, it is slow and it could consume more power because full CMOS swing is needed in certain applications.

Analog approach

The frequency divider with analog approach is also widely used. It tends to be used in the very high frequency millimeter wave range, anywhere from 20 GHz up to 100 GHz. Also, they tend to be in discrete form. The reason for this is the CMOS or BiCMOS process still doesn't have high enough operating frequency range as the more exotic process like pHEMT or heterojunction bipolar transistor (HBT). The analog approach is often called regenerative injection-locked frequency divider (ILFD). Recently, there has been a lot of research to use this technique with CMOS at much lower operating range. However, operating bandwidth is fairly narrow due to the nature of this architecture. In VCO design, one of the key parameters is VCO pulling. Basically, the VCO output frequency will be pulled away if a continuous wave (CW) signal at a different frequency is nearby. VCO pulling is an undesired characteristic, but it is used cleverly to design the ILFD. ILFD is built on this phenomenon by adding a feedback loop, along with a mixer, oscillator or an amplifier with appropriate feedback network and appropriate filters.
From the basic mixer theory, fout=fin-Lo. If fout=Lo, then fin=2*fout. Thus, fin is divided down by half. The active device like an amplifier or a ring oscillator is needed to sustain the oscillation.
The ILFD can be treated as an oscillator operating at fout. The difference is ILFD's output will closely track the input. A good ILFD only oscillates when there is signal injected at the input. The filter is needed to reject all the undesired spurious signals out of the mixer. Otherwise, the ILFD could lock into an undesired frequency output. Since the physical mixer is a non-linear device, it could generate signals at Lo and its harmonics. By carefully controlling which harmonic is generated, higher divider ratio can be achieved. For example, if the Lo's third harmonic is used, then Lo=3*fo. Then fin=3*fo+fo. Thus, the fin=4*fo, a divide-by-4 circuit is accomplished. In an ILFD design, most of the building blocks in a typical receiver are used. Filter, amplifier, oscillator, mixer and feedback network are all rich in theory. There are many good references to cover those topics in details. Some of the references are listed in the reference section. One additional comment with ILFD is on the mixer topology. In the mixer design, harmonics are undesired features. However, as illustrated, the harmonics of the mixer are used to the designer's advantage to design a good ILFD. In the active RFIC mixer implementation, a Gilbert cell double balanced mixer (DBM) is widely used. A well-designed DBM will suppress the even harmonics of the Lo. Then only even divide ratio ILFD can be designed. To design an odd divide ratio ILFD, the even harmonics of the mixer is actually a desired feature. Therefore, a simpler single balanced mixer (SBM) can be used to add more flexibility.

Summary

The frequency divider is an interesting and indispensable building block in RF or microwave circuits. There are two main design approaches: digital logic and analog using injection-lock techniques. The digital logic approach can be further divided as the static DFF and dynamic DFF. Like any other receiver design, design trade offs required depend on the application. In applications from around 40 GHz to 100 GHz, an ILFD is a suitable approach because today's standard CMOS process still doesn't have the bandwidth in that range. From 10 GHz to 40 GHz, successful SCL-type frequency divides have been reported. The power consumption, cost and physical size need to be thoroughly compared to decide if the digital or the analog approach is used. From low GHz to 10 GHz, digital SCL is the working horse in this range. Below the low GHz frequency, a case can be made to use dynamic DFF to reduce the number of the transistors needed, thus the cost and power consumption is reduced as well. In today's RFIC, a programmable divider is often required. The digital approach offers the flexibility and modularity ILFD can't match.
The frequency divider is an actively researched area. There are more clever topologies to be discovered to reduce the size, power consumption, complexity and cost.

Neyker Stewart Zambrano

CRF

http://rfdesign.com/mag/503rfdf1.pdf

Power Amplifiers and Transmitters for RF and Microwave

Power Amplifiers and Transmitters for RF 

and Microwave

I.  INTRODUCTION

A power amplifier (PA) is a circuit for converting dc-input power into a significant amount of RF/microwave output power. In most cases, a PA is not just a small-signal amplifier driven into saturation. There exists a great variety of different PAs, and most employ techniques beyond simple linear amplification. A transmitter contains one or more PAs, as well as ancillary circuits such as signal generators, frequency converters, modulators, signal processors, linearizers, and power supplies. The classic architecture employs progressively larger PAs to boost a low-level signal to the desired output power. However, a wide variety of different architectures in essence disassemble and then reassemble the signal to permit amplification with higher efficiency and linearity.

In the early days of wireless communication (1895-mid-1920s), RF power was generated by spark, arc, and alternator techniques. With the advent of the DeForest audion in 1907, the thermoionic vacuum tube offered a means of generating and controlling RF signals, and vacuum-tube PAs were dominant from the late 1920s through the mid-1970s. Discrete solid-state RF-power devices began to appear at the end of the 1960s with the introduction of silicon bipolar transistors such as the 2N6093 [(75-W HF single sideband (SSB)] by RCA. Their dominance in the 1980s brought about the use of lower voltages, higher currents, and relatively low load resistances. The 1990s saw a proliferation of a variety of new solid-state devices including HEMT, pHEMT, HFET, and HBT, using a variety of new materials such as InP, SiC, and GaN. These devices offer amplification to 100 GHz or more and are in many cases grown to order in MMIC form. The combination of digital signal  processing (DSP) and microprocessor control allows widespread use of complicated feedback and predistortion techniques to improve efficiency and linearity.
Modern applications are highly varied. Frequencies from VLF through millimeter wave (MMW) are used for communication, navigation, and broadcasting. Output powers vary from
10 mW in short-range unlicensed wireless systems to 1 MW in long-range broadcast transmitters. Almost every conceivable type of modulation is being used in one system or another. PAs and transmitters also find use in systems such as radar, RF heating, plasma generation, laser drivers, magnetic-resonance imaging,  and  miniature dc/dc  converters.  No  single  PA  or transmitter technique suits all applications. Many techniques that are now coming into use were devised decades ago, but only recently made possible by advances in signal-processing and control technology.

II. LINEARITY

The need for linearity is one of the principal drivers in the design of modern PAs. Signals such as CW, FM, classical FSK,  and  GMSK (used  in  GSM) have  constant envelopes (amplitudes) and, therefore, do not require linear amplification. Full-carrier amplitude modulation is best produced by high-level amplitude modulation of the final RF PA. Linear amplification is required when the signal contains both amplitude and phase modulation. Examples include SSB voice, television (both NTSC and HDTV), modern shaped-pulse  data modulation (QAM, QPSK, CDMA), and multiple carriers (OFDM).
The requirements for both high data rates and efficient utilization of the increasingly crowded spectrum necessitates the use of shaped data pulses in modern digital signals such as QPSK, QAM, and CDMA. Most systems use raised-cosine shaping, which eliminates intersymbol interference during detection and allows the spectrum to be shaped arbitrarily close to rectangular [1]. This requires the transmission of square-root–raised-cosine (SRRC) data pulses that look much like truncated sinc functions. The resultant modulated carrier  has simultaneous amplitude and phase modulation with a peak-to-average ratio of
3–6 dB.
Applications such as cellular base-stations, satellite repeaters, and active phased arrays require the simultaneous amplification of multiple signals. The signals can, in general, have different amplitudes, different modulations, and irregular frequency spacing. In a number of applications including HF modems and digital broadcasting, it is more convenient to use a large number of carriers with low data rates than a single carrier with a high data rate. Orthogonal frequency division multiplex (OFDM) [2] employs carriers with the same amplitude and modulation, separated in frequency so that modulation products from one carrier are zero at the frequencies of the other carriers. The resultant composite signal has a peak-to-average ratio  in the range of 8–13 dB.

Distortion of the amplified signal can be caused by both amplitude nonlinearity (such as a variable gain) or amplitude-to- phase conversion (produced, for example, by a voltage-variable capacitance). The result is splatter into adjacent channels and impairment of detection. Linearity is characterized, measured, and specified by various techniques, depending upon the specific signal and application.

The carrier-to-intermodulation (C/I) ratio, compares the am- plitude of the desired output carriers to the intermodulation-distortion (IMD) products [3]. Noise-power ratio (NPR) is the ratio of the notch power to the total signal power when a PA is driven by noise with a spectral notch. Adjacent channel power ratio (ACPR) compares the power in an adjacent channel to that of the signal. It is currently the most widely used measure of linearity, but defined differently for each application. Error vector magnitude (EVM) is the distance between the desired and actual signal vectors.

III. EFFICIENCY

Efficiency, like linearity, is a critical factor in PA design. Three definitions of efficiency are commonly used. Drain efficiency is defined as the ratio of RF-output power to dc-input power,

n=P0/Pi, Power-added efficiency (PAE) incorporates the RF-drive power by subtracting it from  the output power, (P0-Pdr)/Pi, PAE gives a reasonable indication of PA performance when gain is high; however, it can become negative for low gains. An overall efficiency such as 

P0/(Pi+Pdr) is usable in all situations. This definition can be varied to include driver dc-input power, the power consumed by supporting circuits, and anything else of interest.

The instantaneous efficiency is the efficiency at one specific output level. For most PAs, the instantaneous efficiency is highest at the peak output power (PEP) and decreases as output decreases. When amplifying signals with time-varying amplitudes, a useful measure of performance is the average efficiency, which is defined [4] as the ratio of the average outuput power to the average dc-input power, navg=P0avg/Piavg.

The probability-density function (PDF) gives the relative amount  of  time  an  envelope  spends  at  various  amplitudes. The PDF of an SRRC signal must generally be determined by simulation or measurement. Multiple carriers produce random-phasor sums and, therefore, have Rayleigh-distributed envelopes. The average input and output powers are found by integrating the product of the variable of interest and the PDF of the envelope over the range of the envelope.

The need to conserve battery power and to avoid interference to other users operating on the same frequency necessitates the transmission of signals whose peak amplitudes are well below the PEP of the transmitter. Since peak power is needed only in the worst-case links, the "backoff" is typically in the range of 10–20 dB. For a single-carrier mobile transmitter, backoff rather than envelope PDF is dominant in determining the average power consumption and average efficiency. The PDF of the transmitting power depends not only upon the distance, but also upon factors such as attenuation by buildings, multipath, and orientation of the mobile antenna [5].

Neyker Stewart Zambrano

CRF

http://nemes.colorado.edu/microwave/papers/2002/MTT_FRpaSCpk_zp_Mar02.pdf

Power Amplifiers and Transmitters for RFand Microwave

Power Amplifiers and Transmitters for RF

and Microwave

I.  INTRODUCTION

A power amplifier (PA) is a circuit for converting dc-input power into a significant amount of RF/microwave output power. In most cases, a PA is not just a small-signal amplifier driven into saturation. There exists a great variety of different PAs, and most employ techniques beyond simple linear amplification. A transmitter contains one or more PAs, as well as ancillary circuits such as signal generators, frequency converters, modulators, signal processors, linearizers, and power supplies. The classic architecture employs progressively larger PAs to boost a low-level signal to the desired output power. However, a wide variety of different architectures in essence disassemble and then reassemble the signal to permit amplification with higher efficiency and linearity.

In the early days of wireless communication (1895-mid-1920s), RF power was generated by spark, arc, and alternator techniques. With the advent of the DeForest audion in 1907, the thermoionic vacuum tube offered a means of generating and controlling RF signals, and vacuum-tube PAs were dominant from the late 1920s through the mid-1970s. Discrete solid-state RF-power devices began to appear at the end of the 1960s with the introduction of silicon bipolar transistors such as the 2N6093 [(75-W HF single sideband (SSB)] by RCA. Their dominance in the 1980s brought about the use of lower voltages, higher currents, and relatively low load resistances. The 1990s saw a proliferation of a variety of new solid-state devices including HEMT, pHEMT, HFET, and HBT, using a variety of new materials such as InP, SiC, and GaN. These devices offer amplification to 100 GHz or more and are in many cases grown to order in MMIC form. The combination of digital signal  processing (DSP) and microprocessor control allows widespread use of complicated feedback and predistortion techniques to improve efficiency and linearity.
Modern applications are highly varied. Frequencies from VLF through millimeter wave (MMW) are used for communication, navigation, and broadcasting. Output powers vary from
10 mW in short-range unlicensed wireless systems to 1 MW in long-range broadcast transmitters. Almost every conceivable type of modulation is being used in one system or another. PAs and transmitters also find use in systems such as radar, RF heating, plasma generation, laser drivers, magnetic-resonance imaging,  and  miniature dc/dc  converters.  No  single  PA  or transmitter technique suits all applications. Many techniques that are now coming into use were devised decades ago, but only recently made possible by advances in signal-processing and control technology.

II. LINEARITY

The need for linearity is one of the principal drivers in the design of modern PAs. Signals such as CW, FM, classical FSK,  and  GMSK (used  in  GSM) have  constant envelopes (amplitudes) and, therefore, do not require linear amplification. Full-carrier amplitude modulation is best produced by high-level amplitude modulation of the final RF PA. Linear amplification is required when the signal contains both amplitude and phase modulation. Examples include SSB voice, television (both NTSC and HDTV), modern shaped-pulse  data modulation (QAM, QPSK, CDMA), and multiple carriers (OFDM).
The requirements for both high data rates and efficient utilization of the increasingly crowded spectrum necessitates the use of shaped data pulses in modern digital signals such as QPSK, QAM, and CDMA. Most systems use raised-cosine shaping, which eliminates intersymbol interference during detection and allows the spectrum to be shaped arbitrarily close to rectangular [1]. This requires the transmission of square-root–raised-cosine (SRRC) data pulses that look much like truncated sinc functions. The resultant modulated carrier  has simultaneous amplitude and phase modulation with a peak-to-average ratio of
3–6 dB.
Applications such as cellular base-stations, satellite repeaters, and active phased arrays require the simultaneous amplification of multiple signals. The signals can, in general, have different amplitudes, different modulations, and irregular frequency spacing. In a number of applications including HF modems and digital broadcasting, it is more convenient to use a large number of carriers with low data rates than a single carrier with a high data rate. Orthogonal frequency division multiplex (OFDM) [2] employs carriers with the same amplitude and modulation, separated in frequency so that modulation products from one carrier are zero at the frequencies of the other carriers. The resultant composite signal has a peak-to-average ratio  in the range of 8–13 dB.

Distortion of the amplified signal can be caused by both amplitude nonlinearity (such as a variable gain) or amplitude-to- phase conversion (produced, for example, by a voltage-variable capacitance). The result is splatter into adjacent channels and impairment of detection. Linearity is characterized, measured, and specified by various techniques, depending upon the specific signal and application.

The carrier-to-intermodulation (C/I) ratio, compares the am- plitude of the desired output carriers to the intermodulation-distortion (IMD) products [3]. Noise-power ratio (NPR) is the ratio of the notch power to the total signal power when a PA is driven by noise with a spectral notch. Adjacent channel power ratio (ACPR) compares the power in an adjacent channel to that of the signal. It is currently the most widely used measure of linearity, but defined differently for each application. Error vector magnitude (EVM) is the distance between the desired and actual signal vectors.

III. EFFICIENCY

Efficiency, like linearity, is a critical factor in PA design. Three definitions of efficiency are commonly used. Drain efficiency is defined as the ratio of RF-output power to dc-input power,

n=P0/Pi, Power-added efficiency (PAE) incorporates the RF-drive power by subtracting it from  the output power, (P0-Pdr)/Pi, PAE gives a reasonable indication of PA performance when gain is high; however, it can become negative for low gains. An overall efficiency such as 

P0/(Pi+Pdr) is usable in all situations. This definition can be varied to include driver dc-input power, the power consumed by supporting circuits, and anything else of interest.

The instantaneous efficiency is the efficiency at one specific output level. For most PAs, the instantaneous efficiency is highest at the peak output power (PEP) and decreases as output decreases. When amplifying signals with time-varying amplitudes, a useful measure of performance is the average efficiency, which is defined [4] as the ratio of the average outuput power to the average dc-input power, navg=P0avg/Piavg.

The probability-density function (PDF) gives the relative amount  of  time  an  envelope  spends  at  various  amplitudes. The PDF of an SRRC signal must generally be determined by simulation or measurement. Multiple carriers produce random-phasor sums and, therefore, have Rayleigh-distributed envelopes. The average input and output powers are found by integrating the product of the variable of interest and the PDF of the envelope over the range of the envelope.

The need to conserve battery power and to avoid interference to other users operating on the same frequency necessitates the transmission of signals whose peak amplitudes are well below the PEP of the transmitter. Since peak power is needed only in the worst-case links, the "backoff" is typically in the range of 10–20 dB. For a single-carrier mobile transmitter, backoff rather than envelope PDF is dominant in determining the average power consumption and average efficiency. The PDF of the transmitting power depends not only upon the distance, but also upon factors such as attenuation by buildings, multipath, and orientation of the mobile antenna [5].

Neyker Stewart Zambrano

CRF

http://nemes.colorado.edu/microwave/papers/2002/MTT_FRpaSCpk_zp_Mar02.pdf

RF and Microwave Power Amplifier and TransmitterTechnologies.Feedback

RF and Microwave Power

Amplifier and TransmitterTechnologies.Feedback

Linearization techniques are incorporated into power amplifiers and transmiters for the dual purposes of improving linearity and for allowing operation with less back-off and therefore higher effi- ciency. This article provides a summary of the three main families of techniques have been developed: Feedback, feedforward, and predistortion.

RF Feedback
The basis of this technique is similar to its audio-frequency counterpart. A portion of the RF-output signal from the amplifier is fed back to, and subtracted from, the RF-input signal without detection or down- conversion. Considerable care must be taken when using feedback at RF as the delays involved must be small to ensure stability. In addition, the loss of gain at RF is generally a more significant sacrifice than it is at audio frequencies. For these reasons, the use of RF feedback in dis- crete circuits is usually restricted to HF and lower VHF frequencies [99]. It can be applied within MMIC devices, however, well into the microwave region.
In an active RF feedback system, the volt- age divider of a conventional passive-feedback system is replaced by an active (amplifier) stage. The gain in the feedback path reduces the power dissipated in the feedback compo- nents. While such systems demonstrate IMD reduction [105], they tend to work best at a specific signal level.

Envelope Feedback

The problem of delay in RF feedback is alleviated to a large extent by utilizing the signal envelope as the feedback parameter. This approach takes care  of in-band distortion products associated  with amplitude  nonlinearity. Harmonic distortion products, which are corrected by RF feedback, are generally not an issue as they can easily be removed by filtering in most applications. Envelope feed- back is therefore a popular and simple technique.Envelope feedback can  be applied to either a complete transmitter or a single power  amplifier. The principles of operation are similar and both are described in detail in [100]. The RF input signal is sampled by a coupler and the envelope of the input sample is detected. The resulting envelope is then fed to one input of a differential amplifier, which subtracts it from a similarly obtained sample of the RF output. The  difference signal, representing the error between the input and output envelopes, is used to drive a modulator in the main RF path. This  modulator modifies the envelope of the  RF signal which drives the RF PA. The envelope of the  resulting output signal is therefore linearized to a degree determined by the loop gain of the feedback process. Examples of this type of system are reported in [101] and  [102].
The degree of linearity improvement that can be obtained when using this technique depends upon the relative levels of the AM-AM and AM-PM conversion in the amplifier. For a VHF BJT  amplifier, AM-AM distortion is dominant and  two-tone IMD is typically reduced by 10 dB. Since AM-PM distortion is not corrected by envelope feedback, no linearity improvement is observed if phase distortion is the dominant form of nonlinearity. This is often the case in, for example, class-C and LDMOS PAs. The use of envelope feedback is therefore generally restricted to relatively linear class-A or AB amplifiers

Polar-Loop Feedback

The polar-loop technique overcomes the fundamental inability of envelope feedback to correct for AM-PM distortion effects [103].  Essentially, a phase-locked loop is added to the envelope feedback system. For a narrowband VHF PA, the improvement in two-tone IMD is typically around 30 dB.The envelope and phase-feedback functions operate essentially independently. In this case, envelope detection occurs at the intermediate frequency (IF), as the input signal  is   assumed to be a modulated carrier at IF. Likewise, phase detection takes place at the IF, with limiting being used to minimize the effects of signal amplitude upon the detected phase. Alternatively, it is possible to supply the envelope and phase modulating signals separately at baseband and to undertake the comparisons there.
The key disadvantage of polar feedback lies in the generally different bandwidths required for  the  amplitude and phase feedback paths.Thus,differing levels of improvement of the AM-AM and AM-PM characteristics usually result, and  this often leads to a poorer overall performance than that achievable from an equivalent Cartesian-loop transmitter. A good example of the difference occurs with a standard two-tone test, which causes the phase-feedback path to cope with a discontinuity at the envelope minima. In general, the phase bandwidth must be five to ten times the envelope bandwidth, which limits available loop gain for a given delay.

Cartesian Feedback

The Cartesian-feedback technique overcomes the problems associated with the wide bandwidth of the signal phase by applying modulation feedback in I and Q(Cartesian) components [104]. Since the I and Q components are the natural outputs of a modern DSP, the Cartesian loop is widely used in PMR and SMR systems.
The basic Cartesian loop  consists of two identical feedback processes operating independently on the I and Q channels. The inputs are applied to differential integrators (in the case of a first-order loop) with the resulting difference (error) signals being modulated onto I and Q subcarriers and up-converted to drive the PA. A sample of the output from the PA is attenuated  and quadrature-down-converted  (synchronously with the up- conversion process). The  resulting quadrature feedback signals then form the second inputs to the input differential  integrators, completing the two feedback loops. The phase shifter shown in the up-converter  local-oscillator path is used to align the phases of the up- and down-conversion processes, thereby ensuring that a negative feed- back system is created and that the phase margin of the system is optimized.The effects of applying Cartesian feedback to a highly nonlinear (class-C) PA  amplifying an IS-136 (DAMPS) signal are shown. The first ACPR is improved by 35 dB and the signal is produced within specifications with an efficiency of 60 percent [100].

Neyker Stewart Zambrano
CRF
http://www.highfrequencyelectronics.com/Archives/Nov03/HFE1103_RaabPart4.pdf

Amplifier and TransmitterTechnologies.Feedback

Linearization techniques are incorporated into power amplifiers and transmiters for the dual purposes of improving linearity and for allowing operation with less back-off and therefore higher effi- ciency. This article provides a summary of the three main families of techniques have been developed: Feedback, feedforward, and predistortion.

RF Feedback
The basis of this technique is similar to its audio-frequency counterpart. A portion of the RF-output signal from the amplifier is fed back to, and subtracted from, the RF-input signal without detection or down- conversion. Considerable care must be taken when using feedback at RF as the delays involved must be small to ensure stability. In addition, the loss of gain at RF is generally a more significant sacrifice than it is at audio frequencies. For these reasons, the use of RF feedback in dis- crete circuits is usually restricted to HF and lower VHF frequencies [99]. It can be applied within MMIC devices, however, well into the microwave region.
In an active RF feedback system, the volt- age divider of a conventional passive-feedback system is replaced by an active (amplifier) stage. The gain in the feedback path reduces the power dissipated in the feedback compo- nents. While such systems demonstrate IMD reduction [105], they tend to work best at a specific signal level.

Envelope Feedback

The problem of delay in RF feedback is alleviated to a large extent by utilizing the signal envelope as the feedback parameter. This approach takes care  of in-band distortion products associated  with amplitude  nonlinearity. Harmonic distortion products, which are corrected by RF feedback, are generally not an issue as they can easily be removed by filtering in most applications. Envelope feed- back is therefore a popular and simple technique.Envelope feedback can  be applied to either a complete transmitter or a single power  amplifier. The principles of operation are similar and both are described in detail in [100]. The RF input signal is sampled by a coupler and the envelope of the input sample is detected. The resulting envelope is then fed to one input of a differential amplifier, which subtracts it from a similarly obtained sample of the RF output. The  difference signal, representing the error between the input and output envelopes, is used to drive a modulator in the main RF path. This  modulator modifies the envelope of the  RF signal which drives the RF PA. The envelope of the  resulting output signal is therefore linearized to a degree determined by the loop gain of the feedback process. Examples of this type of system are reported in [101] and  [102].
The degree of linearity improvement that can be obtained when using this technique depends upon the relative levels of the AM-AM and AM-PM conversion in the amplifier. For a VHF BJT  amplifier, AM-AM distortion is dominant and  two-tone IMD is typically reduced by 10 dB. Since AM-PM distortion is not corrected by envelope feedback, no linearity improvement is observed if phase distortion is the dominant form of nonlinearity. This is often the case in, for example, class-C and LDMOS PAs. The use of envelope feedback is therefore generally restricted to relatively linear class-A or AB amplifiers

Polar-Loop Feedback

The polar-loop technique overcomes the fundamental inability of envelope feedback to correct for AM-PM distortion effects [103].  Essentially, a phase-locked loop is added to the envelope feedback system. For a narrowband VHF PA, the improvement in two-tone IMD is typically around 30 dB.The envelope and phase-feedback functions operate essentially independently. In this case, envelope detection occurs at the intermediate frequency (IF), as the input signal  is   assumed to be a modulated carrier at IF. Likewise, phase detection takes place at the IF, with limiting being used to minimize the effects of signal amplitude upon the detected phase. Alternatively, it is possible to supply the envelope and phase modulating signals separately at baseband and to undertake the comparisons there.
The key disadvantage of polar feedback lies in the generally different bandwidths required for  the  amplitude and phase feedback paths.Thus,differing levels of improvement of the AM-AM and AM-PM characteristics usually result, and  this often leads to a poorer overall performance than that achievable from an equivalent Cartesian-loop transmitter. A good example of the difference occurs with a standard two-tone test, which causes the phase-feedback path to cope with a discontinuity at the envelope minima. In general, the phase bandwidth must be five to ten times the envelope bandwidth, which limits available loop gain for a given delay.

Cartesian Feedback

The Cartesian-feedback technique overcomes the problems associated with the wide bandwidth of the signal phase by applying modulation feedback in I and Q(Cartesian) components [104]. Since the I and Q components are the natural outputs of a modern DSP, the Cartesian loop is widely used in PMR and SMR systems.
The basic Cartesian loop  consists of two identical feedback processes operating independently on the I and Q channels. The inputs are applied to differential integrators (in the case of a first-order loop) with the resulting difference (error) signals being modulated onto I and Q subcarriers and up-converted to drive the PA. A sample of the output from the PA is attenuated  and quadrature-down-converted  (synchronously with the up- conversion process). The  resulting quadrature feedback signals then form the second inputs to the input differential  integrators, completing the two feedback loops. The phase shifter shown in the up-converter  local-oscillator path is used to align the phases of the up- and down-conversion processes, thereby ensuring that a negative feed- back system is created and that the phase margin of the system is optimized.The effects of applying Cartesian feedback to a highly nonlinear (class-C) PA  amplifying an IS-136 (DAMPS) signal are shown. The first ACPR is improved by 35 dB and the signal is produced within specifications with an efficiency of 60 percent [100].

Neyker Stewart Zambrano
CRF
http://www.highfrequencyelectronics.com/Archives/Nov03/HFE1103_RaabPart4.pdf

RF and Microwave Power Amplifier and TransmitterTechnologies. Feedforward and Predistortion.

RF and Microwave Power
Amplifier and TransmitterTechnologies. Feedforward and Predistortion

FEEDFORWARD
The very wide bandwidths (10 to 100 MHz) required in multicarrier applications can render feedback and DSP impractical. In such cases, the feedforward technique can be used to achieve ultralinear operation. In its basic configuration, feedforward typically gives improvements in distortion ranging from 20 to 40 dB.

Operation
In its basic form, a feedforward amplifier consists of two amplifiers (the main and error amplifiers), directional couplers, delay lines and loop control networks [110]. The directional couplers are used for power splitting/combining, and the delay lines ensure operation over a wide bandwidth. Loop-control networks, which consist of amplitude and phase-shifting networks, maintain signal and distortion cancellation within the various feed- forward loops.
The input signal is first split into two paths, with one path going to the high-power main amplifier while the other signal path goes to a delay element. The output signal from the main amplifier contains both the desired signal and distortion. This signal is sampled and scaled using attenuators before being combined with the delayed portion of the input signal, which is regarded as distortion-free. The resulting "error signal" ideally contains only the distortion components in the output of the main amplifier. The error signal is then amplified by the low- power, high-linearity error amplifier, and then combined with a delayed version of the main amplifier output. This second combination ideally cancels the distortion components in the mainamplifier output while leaving the desired signal unaltered.
In practice, there is always some residual desired signal passing through the error amplifier. This is in general not a problem unless the additional power is sufficient in magnitude to degrade the linearity of the error amplifier and hence the linearity of the feedforward transmitter.

Signal Cancellation
Successful isolation of an error signal and the removal of distortion components depend upon precise signal cancellation over a band of frequencies. In practice, cancellation is achieved by the vector addition of signal voltages. The allowable amplitude and phase mismatches for different cancellation levels are shown. For manufactured equipment, realistic values of distortioncancellation are around 25 to 30. The limiting factor is nearly always the bandwidth over which a given accura- cy can be obtained.

Efficiency
The outputs of the main and error amplifiers are typically combined in a directional coupler that both isolates the PAs from each other and provides resistive input impedances. For a typical 10 dB coupling ratio,90 percent of the power from the main PA reaches the output. For the same coupling ratio, only 10 percent of the power from the error amplifier reaches the load, thus the error amplifier must produce ten times the power of the distortion in the main amplifier. The peak-to-average ratio of the error signal is often much higher than that of the desired signal, making amplification of the error signal inherently much less efficient than that of the main signal. As a result, the power consumed by the error amplifier can be a significant fraction (e.g., one third) of that of the main amplifier. In addition, it may be necessary to operate one or both amplifiers well into back-off to improve linearity. The overall average efficiency of a feedforward transmitter may therefore be only 10 to 15 percent for typical multicarrier signals.

Automatic Loop Control
Since feedforward is inherently an open-loop process,
changes in device characteristics over time, temperature, voltage and signal level degrade the amplitude and phase matching and therefore increase distortion in the transmitter output. An automatic control scheme continuously adjusts the gain and phase to achieve the best signal cancellation and output linearity. The first step is to use FFT techniques, direct power measurement, or pilot signals to determine how well the loop is balanced. Both digital and analog techniques can be used for loop control and adjustment. Signal processing can be used to reduce the peaks in multicarrier signals and to keep distortion products out of the nearby receiving band [111].

Performance
An example of the use of feedforward to improve lin- earity is shown in Figure 48. The signal consists of a mix of TDMA and CDMA carriers. The power amplifiers are based upon LDMOS transistors and have two-tone IMD levels in the range –30 to –35 dBc at nominal output power. The addition of feedforward reduces the level of distortion by approximately 30 dB to meet the required levels of better than –60 dBc. The average efficiency is typically about 10 percent.

8c. PREDISTORTION
The basic concept of a predistortion system involves the insertion of a nonlinear element prior to the RF PA such that of both is linear.Predistortion can be accomplished at either RF or the combined transfer characteristic baseband.

RF Predistortion
The block diagram of a simple RF predistorter is shown. A compressive characteristic, created by the nonlinearity in the lower path (e.g., a diode) is subtracted from a linear characteristic(the upper path) to generate an expansive characteristic. The output of the linear path (typically just a time delay)is given by:

vl(vin) = a1vin (1)

and that of the compressive path is given by


invc(vin) = a2vin – bv3 (2)

Subtracting the above equations gives

invpd(vin) = (a2 – a2) vin – bv3 (3)

This  is now an  expansive characteristic with a linear gain  of a1  – a2, and  may  be used to predistort a compres-
sive  amplifier characteristic  (cubic  in  this example)  by
appropriate choice of a1, a2   and b.
An example of the results from using a simple diode-
based RF predistorter with a 120-W LDMOS PA amplifying an IS-95 CDMA signal is shown. When applied to π/4-DQPSK modulation in a satellite application, the same predistorter roughly  halves the EVM, improves the efficiency from 22 to 29 percent, and  doubles the available output power.
Predistortion bandwidths tend to be limited by similar factors to that of feedforward, namely gain and phase flatness of the predistorter itself and of the RF PA. In addition, memory effects in the PA and the predistorter limit the degree cancellation, and these tend to become poorer with increasing bandwidth.
Better performance can be achieved with more complex forms of RF predistortion such as  Adaptive Parametric Linearization (APL®), which is capable of multi-order correction [106]. Most RF-predistortion techniques are capable of broadband operation with practical operational bandwidths similar to, or greater than, those of feedforward.

Digital Predistortion
Digital predistortion techniques exploit the considerable processing power now available from DSP devices, which allows them both to form and to update the required predistortion characteristic. They can operate with analog-baseband, digital-baseband, analog-IF, digital-IF,or analog-RF input signals.Digital-baseband and digital-IF processing are most common.
The two most common types of digital predistorter are termed mapping predistorters [107] and constant-gain predistorters [108]. A mapping predistorter utilizes two look-up tables, each of which is a function of two variables (IIN  and  QIN), as shown . This type of predis- torter  is capable of  excellent  performance. However, it requires a significant storage and/or processing overhead for the look-up tables and their updating mechanism, and has a low speed of convergence. The low convergence speed results from the need to address all points in the I/Q complex plane before convergence can be completed.
 A constant-gain predistorter requires only a single-dimensional look-up table, indexed by the signal envelope. It is therefore a much simpler implementation and requires significantly less memory for a given  level of performance and adaptation time.It uses the look-up table to force the predistorter and associated PA to exhib-it a constant gain and phase at all envelope levels. The overall transfer characteristic is then linear:

GPD(IIN(t),QIN(t))×GPA(IPD(t),QPD(t)) = k     (4)

An example of the improvement in the amplitude-transfer characteristic by an RF- input/output digital predistorter [109]. The plot is based upon real-time using samples from a GSM-EDGE signal. Both the gain expansion and compression are improved by the linearizer. EVM is reduced from around 4.5 to 0.7 percent. The ACPR for IS-136 DAMPS modulation

(π/4- DQPSK) is reduced by nearly 20 dB. When generating mask-compliant EDGE modulation at full out- put power (850-900 MHz), the linearized PA has an efficiency of over 30 percent. An example of linearization of a PA with two 3G W-CDMA signals by a digital baseband-input predistorter. The linearized amplifier meets the required spectral mask with a comfortable margin at all frequency offsets. The noise floor is set by the degree of clipping employed on the waveform, which limits the ACPR improvement obtained. It clearly demonstrates, however, that digital predistortion can be used in broadband as well as narrowband applications.

Neyker Stewart Zambrano

CRF

http://www.highfrequencyelectronics.com/Archives/Nov03/HFE1103_RaabPart4.pdf

RF and Microwave Power Amplifier and Transmitter Technologies

RF and Microwave Power
Amplifier and TransmitterTechnologies

Linearization techniques are incorporated into power amplifiers and transmiters for the dual purposes of improving linearity and for allowing operation with less back-off and therefore higher effi- ciency. This article provides a summary of the three main families of techniques have been developed: Feedback, feedforward, and predistortion.

RF Feedback

The basis of this technique is similar to its audio-frequency counterpart. A portion of the RF-output signal from the amplifier is fed back to, and subtracted from, the RF-input signal without detection or down- conversion. Considerable care must be taken when using feedback at RF as the delays involved must be small to ensure stability. In addition, the loss of gain at RF is generally a more significant sacrifice than it is at audio frequencies. For these reasons, the use of RF feedback in dis- crete circuits is usually restricted to HF and lower VHF frequencies [99]. It can be applied within MMIC devices, however, well into the microwave region.

In an active RF feedback system, the volt- age divider of a conventional passive-feedback system is replaced by an active (amplifier) stage. The gain in the feedback path reduces the power dissipated in the feedback compo- nents. While such systems demonstrate IMD reduction [105], they tend to work best at a specific signal level.

Envelope Feedback
The   problem of  delay   in  RF  feedback  is alleviated to  a  large extent by  utilizing  the signal  envelope as   the   feedback  parameter. This  approach takes care  of in-band distortion products  associated  with  amplitude  nonlin- earity.  Harmonic  distortion  products, which  are   corrected  by  RF  feedback, are   generally not  an  issue as they can  easily be removed by filtering  in  most  applications. Envelope feed- back  is  therefore a  popular and  simple technique.
Envelope feedback can  be applied to either a complete transmitter  or a single power  amplifier. The principles of operation are  similar and  both  are  described in detail in [100]. The RF input signal is sampled by a coupler and the  envelope of the  input sample is  detected. The  resulting envelope is then fed to one input of a differential amplifier, which  subtracts it  from  a similarly

Envelope Feedback
The   problem of  delay   in  RF  feedback  is alleviated to  a  large extent by  utilizing  the signal  envelope as   the   feedback  parameter. This  approach takes care  of in-band distortion products  associated  with  amplitude  nonlin- earity.  Harmonic  distortion  products, which  are   corrected  by  RF  feedback, are   generally not  an  issue as they can  easily be removed by filtering  in  most  applications. Envelope feed- back  is  therefore a  popular and  simple technique.
Envelope feedback can  be applied to either a complete transmitter or a single power  amplifier. The principles of operation are  similar and  both  are  described in detail in [100]. The RF input signal is sam-  pled by a coupler and the  envelope of the  input sample is  detected. The  resulting envelope is then fed to one input of a differential amplifier,   which    subtracts   it  from   a   similarly obtained sample of the  RF  output. The  difference signal, representing  the   error  between the   input  and  output envelopes, is  used to  drive a  modulator  in  the  main RF path. This  modulator modifies the envelope of the  RF sig- nal  which  drives the  RF PA. The envelope of the  resulting output signal is  therefore linearized to  a  degree deter- mined by the loop gain  of the  feedback process. Examples of this type of system are  reported in [101] and  [102].

The   degree  of  linearity  improvement that  can   be obtained when using this technique depends upon the relative levels  of the  AM-AM and  AM-PM conversion in the amplifier. For a VHF  BJT  amplifier, AM-AM distortion is dominant and  two-tone IMD  is  typically reduced by  10 dB. Since  AM-PM  distortion is not  corrected by envelope feedback, no  linearity  improvement is  observed if phase distortion is  the  dominant form  of nonlinearity. This  is often  the  case in, for example, class-C and  LDMOS  PAs. The   use   of   envelope  feedback  is   therefore  generally restricted to relatively linear class-A  or AB amplifiers

Polar-Loop  Feedback
The  polar-loop technique overcomes the  fundamental inability of envelope feedback to correct  for AM-PM  distortion effects   [103].  Essentially, a  phase-locked loop  is added  to   the   envelope  feedback   system  as   shown  in Figure 43. For  a narrowband VHF  PA, the  improvement in two-tone IMD is typically around 30 dB.
The  envelope and  phase-feedback functions  operate essentially independently. In this case, envelope detection occurs  at the  intermediate  frequency (IF),  as  the  input signal  is   assumed  to   be   a   modulated   carrier at  IF. Likewise, phase detection takes place at the  IF, with limiting being  used to minimize the  effects  of signal amplitude upon  the  detected phase. Alternatively, it is possible to supply the  envelope and phase modulating signals separately at baseband  and to  undertake the  comparisons there.

The key disadvantage of polar feedback lies in the generally different bandwidths required for  the  amplitude and   phase  feedback  paths.Thus,differing  levels of improvement of the AM-AM and AM-PM characteristics usually result, and  this often leads to a poorer overall performance than  that achievable from an equivalent Cartesian-loop transmitter. A good example of the difference occurs with a standard two-tone test, which causes the phase-feedback path to cope with a discontinuity at the envelope minima. In general, the phase bandwidth must be five to ten times the envelope bandwidth, which limits available loop gain for a given delay.

Cartesian Feedback
The Cartesian-feedback technique overcomes the problems associated with the wide bandwidth of the signal phase by applying modulation feedback in I and Q(Cartesian) components [104]. Since the I and Q components are the natural outputs of a modern DSP, the Cartesian loop is widely used in PMR and SMR systems.

The  basic  Cartesian loop  consists of  two identical feedback processes operating independently on the  I and  Q channels. The  inputs are applied to differential  integrators (in the case of a first-order loop) with the  resulting difference (error) signals being modulated onto I and  Q subcarriers and up-converted to drive the  PA. A sample of  the output from the PA is  attenuated  and quadrature-down-converted  (synchronously with the up- conversion process). The  resulting quadrature  feedback signals then form the  second inputs to the input differen- tial  integrators, completing the two feedback loops. The phase shifter shown in  the  up-converter  local-oscillator path is used to align the phases of the  up- and down-conversion processes, thereby ensuring that a negative feed- back  system is created and that the  phase margin of the system is optimized.
The  effects  of applying Cartesian feedback to a highly nonlinear  (class-C) PA  amplifying an IS-136 (DAMPS) signal are shown. The first ACPR is improved by 35 dB and the signal is produced within specifications with an efficiency of 60 percent [100].

Neyker Stewart Zambrano

CRF

http://www.highfrequencyelectronics.com/Archives/Nov03/HFE1103_RaabPart4.pdf

EE523 microwave circuit design

EE523
Microwave Circuit Design

· Microwave Circuits are composed of distributed elements with dimensions such that the voltage and phase over the length of the device can vary significantly.
· By modifying the lengths and dimensions of the device, the line voltage (and current) amplitude and phase can be effectively controlled in a manner to obtain a specifically desired frequency response of the device.
· Microwave Circuits are used to design microwave amplifiers, oscillators, filters, power dividers/combiners, multiplexers, antennas and mixers.
· The necessary tools for the analysis and design of microwave circuit devices require an understanding of: transmission lines, two-port networks (Z, Y, ABCD Parameters), network theory
(S-parameters), impedance matching, and filter design. Much of
the course will develop a deeper understanding of these fields while integrating specific applications of microwave circuit design.
· Microwave circuits find their use in a plethora of applications, including:
1) Wireless Communications (cellular, PCS)
2) Wireless Networking
3) Digital Communications (Ground-Ground, Satellite- Ground, Satellite-Satellite)
4) Radar Systems (ground based, airborne, personal vehicles) – Target detection and identification, imaging (e.g., SAR)5) Deep Space Communications
6) Medical Imaging and treatment
7) Radio Spectrometry
What is the Microwave Circuit Designer's Duty for Wireless
Systems?
Designing filters, mixers, amplifiers, oscillators, matching networks, packaging, and system level design of the Analog and Digital Systems
Designing the antennas and matching networks
Propagation Affects (multipath, signal diversity)
What are the principal future challenges?
Miniaturized and low cost microwave circuitry (applies to cellular, PCS, GPS, on-board radar).
Direct hand held unit communication with low earth orbit and mid earth orbit satellites
Direct high-speed digital communication with low earth orbit and mid earth orbit with portable computers.
Miniature high-power amplifiers with low signal to noise ratios, miniature high-Q filters, novel printed antennas, new packaging technology for combining RF and digital circuitry, reducing parasitics, improved modeling and analysis capabilities.

Neyker Stewart Zambrano

CRF

http://www.engr.uky.edu/~gedney/courses/ee523/notes/Set1_Intro.pdf